Fiche technique pour LT3493-3 de Analog Devices Inc.

L7 I [I‘D LT3493-3 TECHNOLOGY L7 5r: 0 0,2 M as m 1.0 I LOAD cumm m) L7LJHMEGB
1
LT3493-3
3493-3f
APPLICATIO S
U
FEATURES
TYPICAL APPLICATIO
U
DESCRIPTIO
U
1.2A, 750kHz Step-Down
Switching Regulator in
2mm × 3mm DFN
The LT
®
3493-3 is a current mode PWM step-down DC/DC
converter with an internal 1.75A power switch. The wide
operating input range of 6.8V to 36V (40V maximum) makes
the LT3493-3 ideal for regulating power from a wide vari-
ety of sources, including unregulated wall transformers,
24V industrial supplies and automotive batteries. Its high
operating frequency allows the use of tiny, low cost induc-
tors and ceramic capacitors, resulting in low, predictable
output ripple.
Cycle-by-cycle current limit provides protection against
shorted outputs and soft-start eliminates input current
surge during start-up. The low current (<2µA) shutdown
mode provides output disconnect, enabling easy power
management in battery-powered systems.
■
Wide Input Range: 6.8V to 36V Operating,
40V Maximum
■
1.2A Output Current
■
Fixed Frequency Operation: 750kHz
■
Output Adjustable Down to 780mV
■
Short-Circuit Robust
■
Uses Tiny Capacitors and Inductors
■
Soft-Start
■
Internally Compensated
■
Low Shutdown Current: <2µA
■
Low V
CESAT
Switch: 330mV at 1A
■
Thermally Enhanced, Low Profile 2mm × 3mm
DFN-6 Package
■
Automotive Battery Regulation
■
Industrial Control Supplies
■
Wall Transformer Regulation
■
Distributed Supply Regulation
■
Battery-Powered Equipment
3.3V Step-Down Converter Efficiency
V
IN
6.8V TO 36V
ON OFF
0.1µF10µH
32.4k
10µF
3493-3 TA01a
22pF
1µF10k
V
OUT
3.3V
1.2A, V
IN
> 12V
1.1A, V
IN
> 8V
V
IN
BOOST
GND FB
SHDN SW
LT3493-3
, LT, LTC and LTM are registered trademarks of Linear Technology Corporation.
ThinSOT is a trademark of Linear Technology Corporation.
All other trademarks are the property of their respective owners.
LOAD CURRENT (A)
EFFICIENCY (%)
70
80
3493-3 TA01b
60
50 0.4 0.8 1.2
0.20 0.6 1.0
90
65
75
55
85
V
IN
= 12V
V
OUT
= 3.3V
L = 10µH
PfiCHflGE/OBDEBI FOB fl'I'IOI'I \ \NCLOGV :2 LT
2
LT3493-3
3493-3f
Input Voltage (V
IN
) .................................................. 40V
BOOST Pin Voltage .................................................. 50V
BOOST Pin Above SW Pin ....................................... 25V
SHDN Pin ................................................................ 40V
FB Voltage ................................................................. 6V
Operating Temperature Range (Note 2)
LT3493E-3........................................... –40°C to 85°C
Maximum Junction Temperature .......................... 125°C
Storage Temperature Range ................. – 65°C to 150°CORDER PART NUMBER DCB PART MARKING
LT3493EDCB-3
ABSOLUTE AXI U RATI GS
WWWU
PACKAGE/ORDER I FOR ATIO
UU
W
(Note 1)
T
JMAX
= 125°C, θ
JA
= 64°C/ W
EXPOSED PAD (PIN 7) IS GND, MUST BE SOLDERED TO PCB
Order Options Tape and Reel: Add #TR Lead Free: Add #PBF
Lead Free Tape and Reel: Add #TRPBF
Lead Free Part Marking: http://www.linear.com/leadfree/
PARAMETER CONDITIONS MIN TYP MAX UNITS
V
IN
Operating Range 6.8 36 V
Undervoltage Lockout 6.2 6.5 6.8 V
Feedback Voltage ●765 780 795 mV
FB Pin Bias Current V
FB
= Measured V
REF
(Note 4) ●50 150 nA
Quiescent Current Not Switching 1.9 2.5 mA
Quiescent Current in Shutdown V
SHDN
= 0V 0.01 2 µA
Reference Line Regulation V
IN
= 6.8V to 36V 0.007 %/V
Switching Frequency V
FB
= 0.7V 685 750 815 kHz
V
FB
= 0V 36 kHz
Maximum Duty Cycle ●88 95 %
T
A
= 25°C9195%
The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C.
VIN = 12V, VBOOST = 17V, unless otherwise noted. (Note 2)
ELECTRICAL CHARACTERISTICS
Consult LTC Marketing for parts specified with wider operating temperature ranges.
TOP VIEW
SHDN
V
IN
SW
FB
GND
BOOST
DCB PACKAGE
6-LEAD (2mm × 3mm) PLASTIC DFN
4
5
7
6
3
2
1
LCGJ
\JOLCGV
3
LT3493-3
3493-3f
PARAMETER CONDITIONS MIN TYP MAX UNITS
Switch Current Limit (Note 3) 1.4 1.75 2.2 A
Switch V
CESAT
I
SW
= 1A 330 mV
Switch Leakage Current 2µA
Minimum Boost Voltage Above Switch I
SW
= 1A 1.85 2.2 V
BOOST Pin Current I
SW
= 1A 30 50 mA
SHDN Input Voltage High 2.3 V
SHDN Input Voltage Low 0.3 V
SHDN Bias Current V
SHDN
= 2.3V (Note 5) 6 15 µA
V
SHDN
= 0V 0.01 0.1 µA
The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C.
VIN = 12V, VBOOST = 17V, unless otherwise noted. (Note 2)
ELECTRICAL CHARACTERISTICS
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LT3493E-3 is guaranteed to meet performance specifications
from 0°C to 85°C. Specifications over the –40°C to 85°C operating
temperature range are assured by design, characterization and correlation
with statistical process controls.
Note 3: Current limit guaranteed by design and/or correlation to static test.
Slope compensation reduces current limit at higher duty cycle.
Note 4: Current flows out of pin.
Note 5: Current flows into pin.
TYPICAL PERFOR A CE CHARACTERISTICS
UW
TA = 25°C unless otherwise noted.
Efficiency (VOUT = 3.3V, L = 10µH)
LOAD CURRENT (A)
0
50
EFFICIENCY (%)
55
65
70
75
0.8 1.0
3493-3 G02
60
0.2 0.4 0.6 1.2
80
85
90
VIN = 8V
VIN = 12V
VIN = 24V
Efficiency (VOUT = 5V, L = 10µH)
LOAD CURRENT (A)
0
50
EFFICIENCY (%)
55
65
70
75
0.8 1.0
95
3493-3 G01
60
0.2 0.4 0.6 1.2
80
85
90
V
IN
= 8V
V
IN
= 12V
V
IN
= 24V
FREQUENCY (kHz) / / TVPmAL / / f / / mm / MIN‘MUM 090 5 ID ‘5 EU 25 3 VwW) / / ago an 5 m 15 2n 25 50 0 5254 ususwnwzuwsm 5m 725 u 25 5o 75 w Vw 1V) SW‘TCH CURRENT (A) “MPEWURE (“0) Switching Frequency Frequency Fnldhack Salt-Stan am sun 20 mm A 700 15 mu ; A I6 5 sun 3’ 74m 5 5 14 E 500 3 72m 3 k 12 a z mu E 400 § In “L 5 BED E 500 2 05 55B S E 06 - 200 54m 5 fi 04 52a um 02 sun 0 0 5m 725 u 25 5a 75 we a me 200 mu ADD 5m euu mu sea 0 025 u5u n V 125‘5n I75 2 TEMPERATURE (a) FEEDBACK VOLTAGE (mvy SHDN PW VOLTAGE (V) 5495 L7L'HWEGB
4
LT3493-3
3493-3f
TYPICAL PERFOR A CE CHARACTERISTICS
UW
Maximum Load Current,
VOUT = 5V, L = 8.2µH
Switch Voltage Drop Undervoltage Lockout
Maximum Load Current,
VOUT = 3.3V, L = 4.7µH
Switching Frequency Frequency Foldback
V
IN
(V)
8
1.60
1.50
1.40
1.30
1.20
1.10
1.00
0.90 20 28
3493-3 G04
12 16 24
OUTPUT CURRENT (A)
TYPICAL
MINIMUM
V
IN
(V)
5
1.40
1.50
1.60
25
3493-3 G05
1.30
1.20
10 15 20 30
1.10
1.00
0.90
OUTPUT CURRENT (A)
TYPICAL
MINIMUM
SWITCH CURRENT (A)
0
V
CE(SW)
(mV)
150
450
500
550
0.4 0.8 1.0
3493-3 G06
50
350
250
100
400
0
300
200
0.2 0.6 1.4
1.2 1.6 1.8
T
A
= 25°C
T
A
= 85°C
T
A
= –40°C
TEMPERATURE (°C)
FREQUENCY (kHz)
720
760
800
3493-3 G09
680
640
700
740
780
660
620
600 –25–50 250 75 100
50
Soft-Start
SHDN PIN VOLTAGE (V)
0
0
SWITCH CURRENT LIMIT (A)
0.2
0.6
0.8
1.0
2.0
1.4
0.50 1 1.25
3493-3 G13
0.4
1.6
1.8
1.2
0.25 0.75 1.50 1.75 2
FEEDBACK VOLTAGE (mV)
0
SWITCHING FREQUENCY (kHz)
400
600
800
3493-3 G11
200
0200 400 600
100 300 500 700
800
300
500
100
700
V
IN
(V)
8
1.60
1.50
1.40
1.30
1.20
1.10
1.00
0.90 20 28
3493-3 G22
12 16 24
OUTPUT CURRENT (A)
TYPICAL
MINIMUM
Maximum Load Current,
VOUT = 5V, L = 33µH
V
IN
(V)
5
1.40
1.50
1.60
25
3493-3 G21
1.30
1.20
10 15 20 30
1.10
1.00
0.90
OUTPUT CURRENT (A)
TYPICAL
MINIMUM
Maximum Load Current,
VOUT = 3.3V, L = 10µH
TA = 25°C unless otherwise noted.
TEMPERATURE (°C)
UVLO (V)
6.6
6.8
7.0
3493-3 G07
6.4
6.2
6.5
6.7
6.9
6.3
6.1
6.0 –25–50 250 75 100
50
SD 23 45 40 (A 35 g H 30 E z 25 E' ‘5 g 20 u m I 15 E ‘3 In E 5 n 0 ‘0 02450000214050020 750725 0 255075100 0 VW (Vt TEMPERATURE (no) Dparating Wavetnrms, Operating Wavalnrms Discnnlinunus Made T . a - _ g. _. _ ~‘ sz i ‘ sz 5V/DW 1 ; ‘ 5wmv .— — .— u-a' —— L- -— 4 ‘LW 0 05va 05mm 0 0 v V001 20.00% W 20me VW=12V Ins/DH] ““1”” vw=12v mymv V001 = 3 av vum = 3 av ‘00 05A \gm=5DmA mow L=VUuH cawmpr Cum=mnF 20 40 so 00 m uuw ova LE (-10) 349:7
5
LT3493-3
3493-3f
TYPICAL PERFOR A CE CHARACTERISTICS
UW
Operating Waveforms
Operating Waveforms,
Discontinuous Mode
SHDN Pin Current
V
SHDN
(V)
0
I
SHDN
(µA)
30
40
50
16
3493-3 G14
20
10
25
35
45
15
5
042 86 12 14 18
10 20
VSW
5V/DIV
IL
0.5A/DIV
0
VOUT
20mV/DIV
VIN = 12V
VOUT = 3.3V
IOUT = 0.5A
L = 10µH
COUT = 10µF
1µs/DIV 3493-3 G19
V
SW
5V/DIV
I
L
0.5A/DIV
0
V
OUT
20mV/DIV
1µs/DIV
3493-3 G20
V
IN
= 12V
V
OUT
= 3.3V
I
OUT
= 50mA
L = 10µH
C
OUT
= 10µF
TA = 25°C unless otherwise noted.
Switch Current Limit
TEMPERATURE (°C)
–50
1.0
SWITCH CURRENT LIMIT (A)
1.1
1.3
1.4
1.5
2.0
1.7
0 25 100
3493-3 G17
1.2
1.8
1.9
1.6
–25 50 75
Switch Current Limit
DUTY CYCLE (%)
0
SWITCH CURRENT LIMIT (A)
1.2
1.6
2.0
80
3493-3 G18
0.8
0.4
1.0
1.4
1.8
0.6
0.2
020 40 60 100
shutdo '_| || || FREUUENEV i FOLDBAEK — — V5 - - u" i] 80 V T T ? n1 L7 JHQB O?
6
LT3493-3
3493-3f
PI FU CTIO S
UUU
FB (Pin 1): The LT3493-3 regulates its feedback pin to
780mV. Connect the feedback resistor divider tap to this
pin. Set the output voltage according to V
OUT
= 0.78V •
(1 + R1/R2). A good value for R2 is 10k.
GND (Pin 2): Tie the GND pin to a local ground plane below
the LT3493-3 and the circuit components. Return the
feedback divider to this pin.
BOOST (Pin 3): The BOOST pin is used to provide a drive
voltage, higher than the input voltage, to the internal
bipolar NPN power switch.
SW (Pin 4): The SW pin is the output of the internal power
switch. Connect this pin to the inductor, catch diode and
boost capacitor.
V
IN
(Pin 5): The V
IN
pin supplies current to the LT3493-3’s
internal regulator and to the internal power switch. This
pin must be locally bypassed.
SHDN (Pin 6): The SHDN pin is used to put the LT3493-
3 in shutdown mode. Tie to ground to shut down the
LT3493-3. Tie to 2.3V or more for normal operation. If the
shutdown feature is not used, tie this pin to the V
IN
pin.
SHDN also provides a soft-start function; see the Applica-
tions Information section.
Exposed Pad (Pin 7): The Exposed Pad must be soldered
to the PCB and electrically connected to ground. Use a
large ground plane and thermal vias to optimize thermal
performance.
BLOCK DIAGRA
W
1
Σ
R
DRIVER Q1
S
OSC
SLOPE
COMP
FREQUENCY
FOLDBACK
INT REG
AND
UVLO
VCgm
780mV
3493-3 BD
2
5
6
Q
Q
3
4
BOOST
SW
FB
R2 R1
VOUT
L1
D2
C3
C1
D1
VIN
C2
VIN
ON OFF
GND
C4
R3
SHDN
n1 met stanis
7
LT3493-3
3493-3f
OPERATIO
U
The LT3493-3 is a constant frequency, current mode step-
down regulator. A 750kHz oscillator enables an RS flip-
flop, turning on the internal 1.75A power switch Q1. An
amplifier and comparator monitor the current flowing
between the V
IN
and SW pins, turning the switch off when
this current reaches a level determined by the voltage at
V
C
. An error amplifier measures the output voltage through
an external resistor divider tied to the FB pin and servos the
V
C
node. If the error amplifier’s output increases, more
current is delivered to the output; if it decreases, less
current is delivered. An active clamp (not shown) on the V
C
node provides current limit. The V
C
node is also clamped
to the voltage on the SHDN pin; soft-start is implemented
by generating a voltage ramp at the SHDN pin using an
external resistor and capacitor.
An internal regulator provides power to the control cir-
cuitry. This regulator includes an undervoltage lockout to
prevent switching when VIN is less than 6.8V. The SHDN
pin is used to place the LT3493-3 in shutdown, discon-
necting the output and reducing the input current to less
than 2µA.
The switch driver operates from either the input or from
the BOOST pin. An external capacitor and diode are used
to generate a voltage at the BOOST pin that is higher than
the input supply. This allows the driver to fully saturate the
internal bipolar NPN power switch for efficient operation.
The oscillator reduces the LT3493-3’s operating frequency
when the voltage at the FB pin is low. This frequency
foldback helps to control the output current during start-
up and overload.
(Refer to Block Diagram)
JLLMJLLUJJUJM: MWWAW L7LJHWW
8
LT3493-3
3493-3f
V
SW
20V/DIV
V
OUT
200mV/DIV
AC COUPLED
C
OUT
= 10µF
V
OUT
= 3V
V
IN
= 30V
I
LOAD
= 0.75A
L = 10µH
2µs/DIV
3493-3 F01
I
L
0.5A/DIV
APPLICATIO S I FOR ATIO
WUUU
FB Resistor Network
The output voltage is programmed with a resistor divider
between the output and the FB pin. Choose the 1%
resistors according to:
RR V
V
OUT
12
078 1=⎛
⎝
⎜⎞
⎠
⎟
.–
R2 should be 20k or less to avoid bias current errors.
Reference designators refer to the Block Diagram.
An optional phase lead capacitor of 22pF between V
OUT
and FB reduces light-load output ripple.
Input Voltage Range
The input voltage range for LT3493-3 applications de-
pends on the output voltage and on the absolute maximum
ratings of the V
IN
and BOOST pins.
The minimum input voltage is determined by either the
LT3493-3’s minimum operating voltage of 6.8V, or by its
maximum duty cycle. The duty cycle is the fraction of time
that the internal switch is on and is determined by the input
and output voltages:
DC VV
VV V
OUT D
IN SW D
=+
+–
where V
D
is the forward voltage drop of the catch diode
(~0.4V) and V
SW
is the voltage drop of the internal switch
(~0.4V at maximum load). This leads to a minimum input
voltage of:
VVV
DC VV
IN MIN OUT D
MAX DSW
()
–=++
with DC
MAX
= 0.91 (0.88 over temperature).
The maximum input voltage is determined by the absolute
maximum ratings of the V
IN
and BOOST pins. For con-
stant-frequency operation, the maximum input voltage is
determined by the minimum duty cycle, DC
MIN
= 0.10. If
the duty cycle requirement is less than DC
MIN
, the part will
enter pulse-skipping mode. The onset of pulse-skipping
occurs at:
VVV
DC VV
IN PS OUT D
MIN DSW()
–=++
In pulse-skipping mode, the part skips pulses to control
the inductor current and regulate the output voltage,
possibly producing a spectrum of frequencies below
750kHz.
Note that this is a restriction on the operating input voltage
to remain in constant-frequency operation; the circuit will
tolerate transient inputs up to the absolute maximum
ratings of the V
IN
and BOOST pins when the output is in
regulation. The input voltage should be limited to V
IN(PS)
during overload conditions (short-circuit or start-up).
Figure 1
mummy; §M\Nww\lw M L7LJHMEGB
9
LT3493-3
3493-3f
Minimum On Time
The part will still regulate the output at input voltages that
exceed V
IN(PS)
(up to 40V), however, the output voltage
ripple increases as the input voltage is increased. Figure 1
illustrates switching waveforms in continuous mode for a
3V output application near V
IN(PS)
= 33V.
As the input voltage is increased, the part is required to
switch for shorter periods of time. Delays associated with
turning off the power switch dictate the minimum on time
of the part. The minimum on time for the LT3493-3 is
130ns. Figure 2 illustrates the switching waveforms when
the input voltage is increased to V
IN
= 35V.
Now the required on time has decreased below the mini-
mum on time of 130ns. Instead of the switch pulse width
becoming narrower to accommodate the lower duty cycle
requirement, the switch pulse width remains fixed at
130ns. In Figure 2 the inductor current ramps up to a value
exceeding the load current and the output ripple increases
to ~200mV. The part then remains off until the output
voltage dips below 100% of the programmed value before
it begins switching again.
Provided that the output remains in regulation and that the
inductor does not saturate, operation above V
IN(PS)
is safe
and will not damage the part. Figure 3 illustrates the
switching waveforms when the input voltage is increased
to its absolute maximum rating of 40V.
APPLICATIO S I FOR ATIO
WUUU
Figure 3
As the input voltage increases, the inductor current ramps
up quicker, the number of skipped pulses increases and
the output voltage ripple increases. For operation above
V
IN(MAX)
the only component requirement is that the
components be adequately rated for operation at the
intended voltage levels.
The part is robust enough to survive prolonged operation
under these conditions as long as the peak inductor
current does not exceed 2.2A. Inductor current saturation
may further limit performance in this operating regime.
Inductor Selection and Maximum Output Current
A good first choice for the inductor value is:
L = 1.6 (V
OUT
+ V
D
)
where V
D
is the voltage drop of the catch diode (~0.4V) and
L is in µH. With this value there will be no subharmonic
oscillation for applications with 50% or greater duty cycle.
The inductor’s RMS current rating must be greater than
your maximum load current and its saturation current
should be about 30% higher. For robust operation in fault
conditions, the saturation current should be above 2.2A.
To keep efficiency high, the series resistance (DCR) should
be less than 0.1Ω. Table 1 lists several vendors and types
that are suitable.
Of course, such a simple design guide will not always
result in the optimum inductor for your application. A
VSW
20V/DIV
VOUT
200mV/DIV
AC COUPLED
COUT = 10µF
VOUT = 3V
VIN = 40V
ILOAD = 0.75A
L = 10µH
2µs/DIV 3493-3 F03
IL
0.5A/DIV
V
SW
20V/DIV
V
OUT
200mV/DIV
AC COUPLED
C
OUT
= 10µF
V
OUT
= 3V
V
IN
= 35V
I
LOAD
= 0.75A
L = 10µH
2µs/DIV
3493-3 F02
I
L
0.5A/DIV
Figure 2
10
LT3493-3
3493-3f
larger value provides a higher maximum load current and
reduces output voltage ripple at the expense of slower
transient response. If your load is lower than 1.2A, then
you can decrease the value of the inductor and operate
with higher ripple current. This allows you to use a
physically smaller inductor, or one with a lower DCR
resulting in higher efficiency. There are several graphs in
the Typical Performance Characteristics section of this
data sheet that show the maximum load current as a
function of input voltage and inductor value for several
popular output voltages. Low inductance may result in
discontinuous mode operation, which is okay, but further
reduces maximum load current. For details of the maxi-
mum output current and discontinuous mode operation,
see Linear Technology Application Note 44.
Catch Diode
Depending on load current, a 1A to 2A Schottky diode is
recommended for the catch diode, D1. The diode must
have a reverse voltage rating equal to or greater than the
maximum input voltage. The ON Semiconductor MBRM140
is a good choice; it is rated for 1A continuous forward
current and a maximum reverse voltage of 40V.
Input Capacitor
Bypass the input of the LT3493-3 circuit with a 1µF or
higher value ceramic capacitor of X7R or X5R type. Y5V
types have poor performance over temperature and ap-
plied voltage and should not be used. A 1µF ceramic is
adequate to bypass the LT3493-3 and will easily handle
the ripple current. However, if the input power source has
high impedance, or there is significant inductance due to
long wires or cables, additional bulk capacitance may be
APPLICATIO S I FOR ATIO
WUUU
Table 1. Inductor Vendors
Vendor URL Part Series Inductance Range (µH) Size (mm)
Sumida www.sumida.com CDRH4D28 1.2 to 4.7 4.5 × 4.5
CDRH5D28 2.5 to 10 5.5 × 5.5
CDRH8D28 2.5 to 33 8.3 × 8.3
Toko www.toko.com A916CY 2 to 12 6.3 × 6.2
D585LC 1.1 to 39 8.1 × 8.0
Würth Elektronik www.we-online.com WE-TPC(M) 1 to 10 4.8 × 4.8
WE-PD2(M) 2.2 to 22 5.2 × 5.8
WE-PD(S) 1 to 27 7.3 × 7.3
necessary. This can be provided with a low performance
electrolytic capacitor.
Step-down regulators draw current from the input supply
in pulses with very fast rise and fall times. The input
capacitor is required to reduce the resulting voltage ripple
at the LT3493-3 and to force this very high frequency
switching current into a tight local loop, minimizing EMI.
A 1µF capacitor is capable of this task, but only if it is
placed close to the LT3493-3 and the catch diode; see the
PCB Layout section. A second precaution regarding the
ceramic input capacitor concerns the maximum input
voltage rating of the LT3493-3. A ceramic input capacitor
combined with trace or cable inductance forms a high
quality (underdamped) tank circuit. If the LT3493-3 circuit
is plugged into a live supply, the input voltage can ring to
twice its nominal value, possibly exceeding the LT3493-
3’s voltage rating. This situation is easily avoided; see the
Hot Plugging Safely section.
Output Capacitor
The output capacitor has two essential functions. Along
with the inductor, it filters the square wave generated by
the LT3493-3 to produce the DC output. In this role it
determines the output ripple so low impedance at the
switching frequency is important. The second function is
to store energy in order to satisfy transient loads and
stabilize the LT3493-3’s control loop.
Ceramic capacitors have very low equivalent series resis-
tance (ESR) and provide the best ripple performance. A
good value is:
C
OUT
= 65/V
OUT
\JOLCGV
11
LT3493-3
3493-3f
APPLICATIO S I FOR ATIO
WUUU
where COUT is in µF. Use X5R or X7R types and keep in
mind that a ceramic capacitor biased with VOUT will have
less than its nominal capacitance. This choice will provide
low output ripple and good transient response. Transient
performance can be improved with a high value capacitor,
but a phase lead capacitor across the feedback resistor R1
may be required to get the full benefit (see the Compen-
sation section).
For small size, the output capacitor can be chosen accord-
ing to:
COUT = 25/VOUT
where COUT is in µF. However, using an output capacitor
this small results in an increased loop crossover fre-
quency and increased sensitivity to noise. A 22pF capaci-
tor connected between VOUT and the FB pin is required to
filter noise at the FB pin and ensure stability.
High performance electrolytic capacitors can be used for
the output capacitor. Low ESR is important, so choose
one that is intended for use in switching regulators. The
ESR should be specified by the supplier and should be
0.1Ω or less. Such a capacitor will be larger than a
ceramic capaci
tor and will have a larger capacitance, be-
cause the capacitor must be large to achieve low ESR.
Table 2 lists several capacitor vendors.
Figure 4 shows the transient response of the LT3493-3
with several output capacitor choices. The output is 3.3V.
The load current is stepped from 250mA to 1A and back to
250mA, and the oscilloscope traces show the output
voltage. The upper photo shows the recommended value.
The second photo shows the improved response (less
voltage drop) resulting from a larger output capacitor
and a phase lead capacitor. The last photo shows the
response to a high performance electrolytic capacitor.
Transient performance is improved due to the large output
capacitance.
BOOST Pin Considerations
Capacitor C3 and diode D2 are used to generate a boost
voltage that is higher than the input voltage. In most cases
a 0.1µF capacitor and fast switching diode (such as the
1N4148 or 1N914) will work well. Figure 5 shows two
ways to arrange the boost circuit. The BOOST pin must be
at least 2.3V above the SW pin for best efficiency. For
outputs of 3.3V and above, the standard circuit (Figure 5a)
is best. For outputs between 3V and 3.3V, use a 0.22µF
capacitor. For outputs between 2.5V and 3V, use a 0.47µF
capacitor and a small Schottky diode (such as the BAT-54).
For lower output voltages the boost diode can be tied to the
input (Figure 5b). The circuit in Figure 5a is more efficient
because the BOOST pin current comes from a lower
voltage source. You must also be sure that the maximum
voltage rating of the BOOST pin is not exceeded.
The minimum operating voltage of an LT3493-3 applica-
tion is limited by the undervoltage lockout (6.8V) and by
the maximum duty cycle as outlined above. For proper
start-up, the minimum input voltage is also limited by the
boost circuit. If the input voltage is ramped slowly, or the
LT3493-3 is turned on with its SHDN pin when the output
Table 2. Capacitor Vendors
Vendor Phone URL Part Series Comments
Panasonic (714) 373-7366 www.panasonic.com Ceramic,
Polymer, EEF Series
Tantalum
Kemet (864) 963-6300 www.kemet.com Ceramic,
Tantalum T494, T495
Sanyo (408) 749-9714 www.sanyovideo.com Ceramic,
Polymer, POSCAP
Tantalum
Murata (404) 436-1300 www.murata.com Ceramic
AVX www.avxcorp.com Ceramic,
Tantalum TPS Series
Taiyo Yuden (864) 963-6300 www.taiyo-yuden.com Ceramic
E 5:4 T753: 3?}: Vaansr * sz ’: Vam MAX Vaaasri Vm 0 vein (5!) Figure 5. Twn Circuits inr Generating the Buns! Wings 1 2 L7 JHEAB #
12
LT3493-3
3493-3f
APPLICATIO S I FOR ATIO
WUUU
Figure 4. Transient Load Response of the LT3493-3 with Different Output Capacitors
as the Load Current is Stepped from 250mA to 1A. VIN = 12V, VOUT = 3.3V, L = 10µH
Figure 5. Two Circuits for Generating the Boost Voltage
V
IN
BOOST
GND
SW
V
IN
LT3493-3
(5a)
D2
V
OUT
C3
V
BOOST
– V
SW
≅ V
OUT
MAX V
BOOST
≅ V
IN
+ V
OUT
3493-3 F05a
VIN
BOOST
GND
SW
VIN
LT3493-3
(5b)
D2
3493-3 F05b
VOUT
C3
VBOOST – VSW ≅ VIN
MAX VBOOST ≅ 2VIN
10µFFB
32.4k
ILOAD
2A/DIV
IL
0.5A/DIV
VOUT
0.1V/DIV
AC COUPLED
ILOAD
2A/DIV
IL
0.5A/DIV
VOUT
0.1V/DIV
AC COUPLED
40µs/DIV
40µs/DIV
ILOAD
2A/DIV
IL
0.5A/DIV
VOUT
0.1V/DIV
AC COUPLED
40µs/DIV
10k
VOUT
3493-3 F04a
3493-3 F04b
3493-3 F04c
FB
VOUT
32.4k
10k
10µF
×2
3.3nF
SANYO
4TPB100M
FB
VOUT
+
32.4k
10k
100µF
..._I W L7LJHWW
13
LT3493-3
3493-3f
RUN
V
SW
10V/DIV
V
IN
= 12V
V
OUT
= 3.3V
L = 10µH
C
OUT
= 10µF
V
OUT
2V/DIV
20µs/DIV
I
L
0.5A/DIV
V
SW
10V/DIV
V
IN
= 12V
V
OUT
= 3.3V
L = 10µH
C
OUT
= 10µF
V
OUT
2V/DIV
20µs/DIV
I
L
0.5A/DIV
SHDN
GND
3493-3 F07a
RUN
15k
0.1µF
SHDN
GND
3493-3 F07b
is already in regulation, then the boost capacitor may not
be fully charged. Because the boost capacitor is charged
with the energy stored in the inductor, the circuit will rely
on some minimum load current to get the boost circuit
running properly. This minimum load will depend on the
input and output voltages, and on the arrangement of the
boost circuit. The minimum load generally goes to zero
once the circuit has started. In many cases the discharged
output capacitor will present a load to the switcher which
will allow it to start. For 3.3V applications, the undervolt-
age lockout is high enough (6.8V) that the boost capacitor
always gets charged. For 5V applications with output
loads less than 20mA, the minimum input voltage required
to charge the boost capacitor is 6.8V. Note this higher
input voltage requirements is only in worst-case situation
where V
IN
is being ramped very slowly. For lower start-up
voltage, the boost diode can be tied to V
IN
; however this
APPLICATIO S I FOR ATIO
WUUU
restricts the input range to one-half of the absolute maxi-
mum rating of the BOOST pin.
At light loads, the inductor current becomes discontinu-
ous and the effective duty cycle can be very high. This
reduces the minimum input voltage to approximately
400mV above V
OUT
. At higher load currents, the inductor
current is continuous and the duty cycle is limited by the
maximum duty cycle of the LT3493-3, requiring a higher
input voltage to maintain regulation.
Soft-Start
The SHDN pin can be used to soft-start the LT3493-3,
reducing the maximum input current during start-up. The
SHDN pin is driven through an external RC filter to create
a voltage ramp at this pin. Figure 6 shows the start-up
waveforms with and without the soft-start circuit. By
Figure 6. To Soft-Start the LT3493-3, Add a Resistor and Capacitor to the SHDN pin. VIN = 12V, VOUT = 3.3V, COUT = 10µF, RLOAD = 5Ω
if“; T 2T1? 2: DA Prevents a Shnned Inpullmm Discharging Tied In the Output; It Also Protects the circuit Input. The LT3493-3 Runs Only When the Input L7TELQB
14
LT3493-3
3493-3f
choosing a large RC time constant, the peak start up
current can be reduced to the current that is required to
regulate the output, with no overshoot. Choose the value
of the resistor so that it can supply 20µA when the SHDN
pin reaches 2.3V.
Shorted and Reversed Input Protection
If the inductor is chosen so that it won’t saturate exces-
sively, an LT3493-3 buck regulator will tolerate a shorted
output. There is another situation to consider in systems
where the output will be held high when the input to the
LT3493-3 is absent. This may occur in battery charging
applications or in battery backup systems where a battery
or some other supply is diode OR-ed with the LT3493-3’s
output. If the V
IN
pin is allowed to float and the SHDN pin
is held high (either by a logic signal or because it is tied to
V
IN
), then the LT3493-3’s internal circuitry will pull its
quiescent current through its SW pin. This is fine if your
system can tolerate a few mA in this state. If you ground
the SHDN pin, the SW pin current will drop to essentially
zero. However, if the V
IN
pin is grounded while the output
is held high, then parasitic diodes inside the LT3493-3 can
pull large currents from the output through the SW pin and
the V
IN
pin. Figure 7 shows a circuit that will run only when
the input voltage is present and that protects against a
shorted or reversed input.
Hot Plugging Safely
The small size, robustness and low impedance of ceramic
capacitors make them an attractive option for the input
bypass capacitor of LT3493-3 circuits. However, these
capacitors can cause problems if the LT3493-3 is plugged
into a live supply (see Linear Technology Application Note
88 for a complete discussion). The low loss ceramic
capacitor combined with stray inductance in series with
the power source forms an underdamped tank circuit, and
the voltage at the V
IN
pin of the LT3493-3 can ring to twice
the nominal input voltage, possibly exceeding the LT3493-
3’s rating and damaging the part. If the input supply is
poorly controlled or the user will be plugging the LT3493-
3 into an energized supply, the input network should be
designed to prevent this overshoot.
Figure 8 shows the waveforms that result when an LT3493-
3 circuit is connected to a 24V supply through six feet of
24-gauge twisted pair. The first plot is the response with
a 2.2µF ceramic capacitor at the input. The input voltage
rings as high as 35V and the input current peaks at 20A.
One method of damping the tank circuit is to add another
capacitor with a series resistor to the circuit. In Figure 8b
an aluminum electrolytic capacitor has been added. This
capacitor’s high equivalent series resistance damps the
circuit and eliminates the voltage overshoot. The extra
capacitor improves low frequency ripple filtering and can
slightly improve the efficiency of the circuit, though it is
likely to be the largest component in the circuit. An
alternative solution is shown in Figure 8c. A 1Ω resistor is
added in series with the input to eliminate the voltage
overshoot (it also reduces the peak input current). A 0.1µF
capacitor improves high frequency filtering. This solution
is smaller and less expensive than the electrolytic capaci-
tor. For high input voltages its impact on efficiency is
minor, reducing efficiency less than one half percent for a
5V output at full load operating from 24V.
APPLICATIO S I FOR ATIO
WUUU
Figure 7. Diode D4 Prevents a Shorted Input from Discharging
a Backup Battery Tied to the Output; It Also Protects the Circuit
from a Reversed Input. The LT3493-3 Runs Only When the Input
is Present
V
IN
BOOST
GND FB
SHDN SW
D4
V
IN
LT3493-3
3493-3 F08
V
OUT
BACKUP
{H L7LJHMEGB
15
LT3493-3
3493-3f
APPLICATIO S I FOR ATIO
WUUU
Figure 8. A Well Chosen Input Network Prevents Input Voltage Overshoot and
Ensures Reliable Operation When the LT3493-3 is Connected to a Live Supply
Frequency Compensation
The LT3493-3 uses current mode control to regulate the
output. This simplifies loop compensation. In particular,
the LT3493-3 does not require the ESR of the output
capacitor for stability allowing the use of ceramic capaci-
tors to achieve low output ripple and small circuit size.
Figure 9 shows an equivalent circuit for the LT3493-3
control loop. The error amp is a transconductance ampli-
fier with finite output impedance. The power section,
consisting of the modulator, power switch and inductor, is
modeled as a transconductance amplifier generating an
output current proportional to the voltage at the V
C
node.
Note that the output capacitor integrates this current, and
that the capacitor on the V
C
node (C
C
) integrates the error
amplifier output current, resulting in two poles in the loop.
R
C
provides a zero. With the recommended output capaci-
tor, the loop crossover occurs above the R
C
C
C
zero. This
simple model works well as long as the value of the
+
+
LT3493-3
2.2µF
V
IN
20V/DIV
I
IN
5A/DIV
20µs/DIV
V
IN
CLOSING SWITCH
SIMULATES HOT PLUG
I
IN
(8a)
(8b)
(8c)
LOW
IMPEDANCE
ENERGIZED
24V SUPPLY
STRAY
INDUCTANCE
DUE TO 6 FEET
(2 METERS) OF
TWISTED PAIR
+
+
LT3493-3
2.2µF
10µF
35V
AI.EI.
LT3493-3
2.2µF0.1µF
1Ω
3493-3 F09
V
IN
20V/DIV
I
IN
5A/DIV
20µs/DIV
V
IN
20V/DIV
I
IN
5A/DIV
20µs/DIV
DANGER!
RINGING V
IN
MAY EXCEED
ABSOLUTE MAXIMUM
RATING OF THE LT3493-3
–
+
–
+
780mV
SW
VC
LT3493-3
GND
3493-3 F10
R1
OUT
ESR
ERROR
AMPLIFIER
CURRENT MODE
POWER STAGE
FB
R2
1M
RC
60k
CC
100pF
C1
C1
gm =
300µA/V
gm =
1.6A/V
+
CPL
0.7V
Figure 9. Model for Loop Response
inductor is not too high and the loop crossover frequency
is much lower than the switching frequency. With a larger
ceramic capacitor (very low ESR), crossover may be lower
and a phase lead capacitor (C
PL
) across the feedback
divider may improve the phase margin and transient
uuuuuu m \NCLOGV :
16
LT3493-3
3493-3f
APPLICATIO S I FOR ATIO
WUUU
response. Large electrolytic capacitors may have an ESR
large enough to create an additional zero, and the phase
lead may not be necessary.
If the output capacitor is different than the recommended
capacitor, stability should be checked across all operating
conditions, including load current, input voltage and tem-
perature. The LT1375 data sheet contains a more thor-
ough discussion of loop compensation and describes how
to test the stability using a transient load.
PCB Layout
For proper operation and minimum EMI, care must be
taken during printed circuit board layout. Figure 11 shows
the recommended component placement with trace,
ground plane and via locations. Note that large, switched
currents flow in the LT3493-3’s V
IN
and SW pins, the catch
diode (D1) and the input capacitor (C2). The loop formed
by these components should be as small as possible and
tied to system ground in only one place. These compo-
nents, along with the inductor and output capacitor,
should be placed on the same side of the circuit board, and
their connections should be made on that layer. Place a
local, unbroken ground plane below these components,
and tie this ground plane to system ground at one location,
ideally at the ground terminal of the output capacitor C1.
The SW and BOOST nodes should be as small as possible.
Finally, keep the FB node small so that the ground pin and
ground traces will shield it from the SW and BOOST nodes.
Include vias near the exposed GND pad of the LT3493-3 to
help remove heat from the LT3493-3 to the ground plane.
High Temperature Considerations
The die temperature of the LT3493-3 must be lower than
the maximum rating of 125°C. This is generally not a
concern unless the ambient temperature is above 85°C.
For higher temperatures, care should be taken in the
layout of the circuit to ensure good heat sinking of the
LT3493-3. The maximum load current should be derated
as the ambient temperature approaches 125°C. The die
temperature is calculated by multiplying the LT3493-3
power dissipation by the thermal resistance from junction
to ambient. Power dissipation within the LT3493-3 can be
estimated by calculating the total power loss from an
efficiency measurement and subtracting the catch diode
C2 D1 C1
SYSTEM
GROUND
: VIAS TO LOCAL GROUND PLANE
: OUTLINE OF LOCAL GROUND PLANE
V
OUT
3493-3 F11
V
IN
SHUTDOWN
Figure 10. A Good PCB Layout Ensures Proper, Low EMI Operation
i 335W 1.8V Step-Dawn Converter a? , if”? —— 2 M
17
LT3493-3
3493-3f
APPLICATIO S I FOR ATIO
WUUU
0.78V Step-Down Converter
1.8V Step-Down Converter
TYPICAL APPLICATIO S
U
V
IN
6.8V TO 25V
0.1µF3.3µH
MBRM140
47µF
3493-3 TA02
2.2µF
V
OUT
0.78V
1.2A
V
IN
BOOST
GND FB
SHDN SW
LT3493-3
1N4148
ON OFF
VIN
6.8V TO 25V
0.1µF5µH
MBRM140 26.1k
22µF
3493-3 TA03
2.2µF20k
VOUT
1.8V
1.2A
VIN BOOST
GND FB
SHDN SW
LT3493-3
1N4148
ON OFF
loss. The resulting temperature rise at full load is nearly
independent of input voltage. Thermal resistance de-
pends on the layout of the circuit board, but 64°C/W is
typical for the (2mm × 3mm) DFN (DCB) package.
Outputs Greater Than 6V
For outputs greater than 6V, add a resistor of 1k to 2.5k
across the inductor to damp the discontinuous ringing of
the SW node, preventing unintended SW current. The 12V
Step-Down Converter circuit in the Typical Applications
section shows the location of this resistor. Also note that
for outputs above 6V, the input voltage range will be
limited by the maximum rating of the BOOST pin. The 12V
circuit shows how to overcome this limitation using an
additional zener diode.
Other Linear Technology Publications
Application notes AN19, AN35 and AN44 contain more
detailed descriptions and design information for Buck
regulators and other switching regulators. The LT1376
data sheet has a more extensive discussion of output
ripple, loop compensation and stability testing. Design
Note DN100 shows how to generate a bipolar output
supply using a Buck regulator.
3.3V Step-D Vw 5 av m 35v 0N OFF BOOST Lmsaa SHDN Vm sw FB 5v Step-D Vw 5 av T0 35v 0N OFF EnosT Lmsara SHDN Vm sw FB 1 8 L7L|HWEI§Q
18
LT3493-3
3493-3f
3.3V Step-Down Converter
5V Step-Down Converter
TYPICAL APPLICATIO S
U
V
IN
6.8V TO 36V
0.1µF8.2µH
1N4148
MBRM140 32.4k
10µF
3493-3 TA05
1µF10k
V
OUT
3.3V
1A, V
IN
> 7V
1.2A, V
IN
> 12V
V
IN
BOOST
GND FB
SHDN SW
LT3493-3
ON OFF
V
IN
6.8V TO 36V
0.1µF10µH
1N4148
MBRM140 59k
10µF
3493-3 TA06
1µF11k
V
OUT
5V
0.9A, V
IN
> 7V
1.1A, V
IN
> 14V
V
IN
BOOST
GND FB
SHDN SW
LT3493-3
ON OFF
2.5V Step-Down Converter
VIN
6.8V TO 28V
0.47µF6.8µH
BAT54
MBRM140 22.1k
22µF
3493-3 TA04
1µF10k
VOUT
2.5V
1A, VIN > 7V
1.2A, VIN > 10V
VIN BOOST
GND FB
SHDN SW
LT3493-3
ON OFF
I I I6510 057} (2SIDES) I’ 2T5:O as l I I 355:005 135 to D5 (2 SIDES) RECOMMENDED SDLDER PAD PITCH AND ZUUflHD I" (2 SIDES) 4‘ PIN I BAR TDP MARK (SEE NOTE 5)}— D 200 REE I 1 WW I I DRAWING TD BE MADE AJEDEC PA ALL DIMENSIONS ARE IN MILLIME DIMENSIONS DE EXPOSED PAD ON MOLD ELASH MOLD FLASH‘ IE PRE 5 EXPOSED PAD SHALL BE SOLDER P N I 2 DRAWING MDT TD SCALE 3 4 5 SHADED AREA IS ONLY A REEEREN TOP AND BOTTOM DE PACKAGE Imavmuan Nmusm by LIneaT Tacnnmogy Cm Hawever ITO IESparISIhIIIIyIsaSSumBOlnTIIs use L IaIIanmaME mImunnecImumImmmmIsasaesm
19
LT3493-3
3493-3f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-
tation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
PACKAGE DESCRIPTION
U
DCB Package
6-Lead Plastic DFN (2mm × 3mm)
(Reference LTC DWG # 05-08-1715)
3.00 ±0.10
(2 SIDES)
2.00 ±0.10
(2 SIDES)
NOTE:
1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (TBD)
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE
TOP AND BOTTOM OF PACKAGE
0.40 ± 0.10
BOTTOM VIEW—EXPOSED PAD
1.65 ± 0.10
(2 SIDES)
0.75 ±0.05
R = 0.115
TYP
R = 0.05
TYP
1.35 ±0.10
(2 SIDES)
1
3
64
PIN 1 BAR
TOP MARK
(SEE NOTE 6)
0.200 REF
0.00 – 0.05
(DCB6) DFN 0405
0.25 ± 0.05
0.50 BSC
PIN 1 NOTCH
R0.20 OR 0.25
× 45° CHAMFER
0.25 ± 0.05
1.35 ±0.05
(2 SIDES)
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
1.65 ±0.05
(2 SIDES)
2.15 ±0.05
0.70 ±0.05
3.55 ±0.05
PACKAGE
OUTLINE
0.50 BSC
HlF 2|. ‘ron cominuous OF USE 'rwo 2k 025w Di cMDzszasB BELflTGD PflflTS PART NUMBER DESCRIPTION LTWGS 6W1 2A imp EDUKHZ‘ Hign Eiiiciency SieprDown DC/DC Convener LT1767 25V‘ 1 2A mm 1 ZMHZ, High Efficiency SieorDown DC/DC Convener [[1933 SDDmA iom, 500kHz SieorDown Swnching Regulator in SOTVZS LTTBSS 36V 1 4A imp EDUKHZ‘ Hign Eiiiciency SieprDown DC/DC Convener LTTBAD Dual 25V, 1.4A Iour,1 iMHz‘ High Eiliciency SleprDow DC/DC Convener LT197S/LT1977 6W1 2A imp ZDOKHZ m SOUkHz‘ High Eiliciency Sle DC/DC Convener vvim Burst Modegfloeraiion LT3434/LT3435 SW 2 4A imp ZDOKHZ m SOUkHz‘ High Eiliciency Sle L7H” EAR CLOGY
20
LT3493-3
3493-3f
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900
●
FAX: (408) 434-0507
●
www.linear.com
© LINEAR TECHNOLOGY CORPORATION 2006
PART NUMBER DESCRIPTION COMMENTS
LT1766 60V, 1.2A I
OUT
, 200kHz, High Efficiency Step-Down V
IN
: 5.5V to 60V, V
OUT(MIN)
= 1.2V, I
Q
= 2.5mA, I
SD
= 25µA,
DC/DC Converter TSSOP16 and TSSOP16E Packages
LT1767 25V, 1.2A I
OUT
, 1.2MHz, High Efficiency Step-Down V
IN
: 3V to 25V, V
OUT(MIN)
= 1.2V, I
Q
= 1mA, I
SD
= 6µA,
DC/DC Converter MS8E Package
LT1933 500mA I
OUT
, 500kHz Step-Down Switching Regulator V
IN
: 3.6V to 36V, V
OUT(MIN)
= 1.2V, I
Q
= 1.6mA, I
SD
= 1µA,
in SOT-23 ThinSOT Package
LT1936 36V, 1.4A I
OUT
, 500kHz, High Efficiency Step-Down V
IN
: 3.6V to 36V, V
OUT(MIN)
= 1.2V, I
Q
= 1.9mA, I
SD
= 1µA,
DC/DC Converter MS8E Package
LT1940 Dual 25V, 1.4A I
OUT
, 1.1MHz, High Efficiency Step-Down V
IN
: 3.6V to 25V, V
OUT(MIN)
= 1.2V, I
Q
= 3.5mA, I
SD
= <30µA,
DC/DC Converter TSSOP16E Package
LT1976/LT1977 60V, 1.2A I
OUT
, 200kHz to 500kHz, High Efficiency Step-Down V
IN
: 3.3V to 60V, V
OUT(MIN)
= 1.2V, I
Q
= 100µA, I
SD
= <1µA,
DC/DC Converter with Burst Mode® Operation TSSOP16E Package
LT3434/LT3435 60V, 2.4A I
OUT
, 200kHz to 500kHz, High Efficiency Step-Down V
IN
: 3.3V to 60V, V
OUT(MIN)
= 1.2V, I
Q
= 100µA, I
SD
= <1µA,
DC/DC Converter with Burst Mode Operation TSSOP16E Package
LT3437 60V, 400mA I
OUT
, MicroPower Step-Down DC/DC Converter V
IN
: 3.3V to 60V, V
OUT(MIN)
= 1.25V, I
Q
= 100µA, I
SD
= <1µA,
with Burst Mode Operation 3mm × 3mm DFN-10 and TSSOP16E Packages
LT3481 34V, 2A I
OUT
, 3MHz, MicroPower Step-Down DC/DC V
IN
: 3.3V to 34V, V
OUT(MIN)
= 1.265V, I
Q
= 50µA, I
SD
= 1µA,
Converter with I
Q
= 50µA 3mm × 3mm DFN-10 and MS10E Packages
LT3493 36V, 1.4A I
OUT
, 750MHz, High Efficiency Step-Down V
IN
: 3.6V to 36V, V
OUT(MIN)
= 0.8V, I
Q
= 1.9mA, I
SD
= 1µA,
DC/DC Converter 2mm × 3mm DFN Package
LT3505 36V, 1.2A I
OUT
, 750kHz, High Efficiency Step-Down V
IN
: 3.6V to 36V, V
OUT(MIN)
= 0.8V, I
Q
= 1.9mA, I
SD
= 1µA,
DC/DC Converter 3mm × 3mm DFN-8 and MS8E Packages
LT3506/LT3506A Dual 25V, 1.6A I
OUT
, 575kHz to 1.1MHz, High Efficiency V
IN
: 3.6V to 25V, V
OUT(MIN)
= 0.8V, I
Q
= 3.8mA, I
SD
= <30µA,
Step-Down DC/DC Converter 4mm × 5mm DFN-16 Package
Burst Mode is a registered trademark of Linear Technology Corporation.
RELATED PARTS
12V Step-Down Converter
LT 0906 • PRINTED IN USA
TYPICAL APPLICATIO
U
V
IN
14.5V TO 36V
0.1µF22µH
1k*
0.25W
*FOR CONTINUOUS OPERATION ABOVE 30V
USE TWO 2k, 0.25W RESISTORS IN PARALLEL.
D1: CMDZ5235B
1N4148
D1
6V
MBRM140 71.5k
4.7µF
3493-3 TA07
1µF4.99k
V
OUT
12V
1A
V
IN
BOOST
GND FB
SHDN SW
LT3493-3
ON OFF